1. Field of the Invention
The present invention relates in general to the field of signal processing, and more specifically to current sensing in a switching power converter.
2. Description of the Related Art
Power control systems often utilize a switching power converter to convert alternating current (AC) voltages to direct current (DC) voltages or DC-to-DC. Switching power converters often include a nonlinear energy transfer process to provide power factor corrected energy to a load. Power control systems provide power factor corrected and regulated output voltages to many devices that utilize a regulated output voltage.
FIG. 1 represents a power control system 100, which includes a switching power converter 102. Voltage source 101 supplies an alternating current (AC) input voltage Vin to a full bridge diode rectifier 103. The voltage source 101 is, for example, a public utility, and the AC voltage Vin is, for example, a 60 Hz/110 V line voltage in the United States of America or a 50 Hz/220 V line voltage in Europe. The rectifier 103 rectifies the input voltage Vin and supplies a rectified, time-varying, line input voltage VX to the switching power converter 102.
The power control system 100 includes a PFC and output voltage controller 114 to control power factor correction and regulate an output voltage VC of switching power converter 102. Switch 108 is a control switch. The PFC and output voltage controller 114 controls an ON (i.e. conductive) and OFF (i.e. nonconductive) state of switch 108 by varying a state of pulse width modulated control signal CS0. Switching between states of switch 108 regulates the transfer of energy from the rectified line input voltage VX through inductor 110 to capacitor 106. The inductor current iL ramps ‘up’ when the switch 108 conducts, i.e. is “ON”. The inductor current iL ramps down when switch 108 is nonconductive, i.e. is “OFF”, and supplies current iL to recharge capacitor 106. The time period during which inductor current iL ramps down is commonly referred to as the “inductor flyback time”. During the inductor flyback time, boost diode 111 is forward biased. Diode 111 prevents reverse current flow into inductor 110 when switch 108 is OFF. In at least one embodiment, the switching power converter 102 operates in discontinuous current mode, i.e. the inductor current iL ramp up time plus the inductor flyback time is less than the period of the control signal CS0, which controls the conductivity of switch 108.
When switching power converter 102 operates in discontinuous conduction mode, input current iL is proportionate to the ‘on-time’ of switch 108, and the energy transferred to inductor 110 is proportionate to the ‘on-time’ squared. Thus, the energy transfer process is one embodiment of a nonlinear process. In at least one embodiment, control signal CS0 is a pulse width modulated signal, and the switch 108 is a field effect transistor (FET), such as an n-channel FET. Control signal CS0 is a gate voltage of switch 108, and switch 108 conducts when the pulse width of CS0 is high. Thus, the ‘on-time’ of switch 108 is determined by the pulse width of control signal CS0. Accordingly, the energy transferred to inductor 110 is proportionate to a square of the pulse width of control signal CS0.
Capacitor 106 supplies stored energy to load 112. The capacitor 106 is sufficiently large so as to maintain a substantially constant output voltage VC, as established by PFC and output voltage controller 114. The output voltage VC remains substantially constant during constant load conditions. However, as load conditions change, the output voltage VC changes. The PFC and output voltage controller 114 responds to the changes in VC and adjusts the control signal CS0 to restore a substantially constant output voltage as quickly as possible. The switching power converter 102 includes a small capacitor 115 to filter any high frequency signals from the line input voltage VX.
The PFC and output voltage controller 114 controls power factor correction of switching power converter 102 and an amount of energy transferred to load 112. The goal of power factor correction technology is to make the switching power converter 102 appear resistive to the voltage source 101. Thus, PFC and output voltage controller 114 attempts to control the inductor current iL so that the average inductor current iL is linearly and directly related to the line input voltage VX. The PFC and output voltage controller 114 controls the pulse width (PW) and period (TT) of control signal CS0 so that a desired amount of energy is transferred to capacitor 106. The desired amount of energy depends upon the voltage and current requirements of load 112.
To regulate the amount of energy transferred and maintain a power factor close to one, PFC and output voltage controller 114 varies the period of control signal CS0 so that the input current iL tracks the changes in input voltage VX and holds the output voltage VC constant. Thus, as the input voltage VX increases, PFC and output voltage controller 114 increases the period TT of control signal CS0, and as the input voltage VX decreases, PFC and output voltage controller 114 decreases the period of control signal CS0. At the same time, the pulse width PW of control signal CS0 is adjusted to maintain a constant duty cycle (D) of control signal CS0, and, thus, hold the output voltage VC constant. In at least one embodiment, the PFC and output voltage controller 114 updates the control signal CS0 at a frequency much greater than the frequency of input voltage VX. The frequency of input voltage VX is generally 50-60 Hz. The frequency 1/TT of control signal CS0 is, for example, between 20 kHz and 130 kHz. Frequencies at or above 20 kHz avoid audio frequencies and frequencies at or below 130 kHz avoid significant switching inefficiencies while still maintaining good power factor, e.g. between 0.9 and 1, and an approximately constant output voltage VC.
In addition to sensing input voltage VX and output voltage VC, PFC and output voltage controller 114 also senses current iRsense—0 across current sense resistor 116. Current sense resistor 116 is connected to switch 108 and rectifier 103 on an input side of power control system 100. PFC and output voltage controller 114 senses current iRsense—0 by sensing the voltage across current sense resistor 116 and determining the sense current iRsense—0 from the sensed voltage and the known value of sense resistor 116.
Referring to FIG. 2, signal graphs 200 depict the relationship between sense current iRsense—0 and control signal CS0 for a high root mean square (RMS) input voltage VX—HIGH RMS and a low voltage input voltage VX—LOW RMS. Signal graphs 200 depict three exemplary periods T(0), T(1), and T(2) of control signal CS0 and sense current iRsense—0. In at least one embodiment, the time marks t0 through t9 mark identical time for FIGS. 2, 4, and 6 for comparison purposes. The input voltage VX can vary by a few volts due to slight load changes or other causes or vary by at least tens of volts due to, for example, dramatic surges in power demand. The input voltage VX can also vary due to, for example, traveling from a country with a 110 V nominal line input voltage Vin to a country with a 220 V nominal line input voltage Vin. The sense resistor 116 is sized to produce a measurable signal for both a high RMS input voltage VX—HIGH RMS and a low voltage input voltage VX—LOW RMS.
In general, when control signal CS0 is high, switch 108 conducts (“ON”) and inductor current iL flows through both switch 108 and current sense resistor 116. The sense current iRsense—0 tracks the inductor current iL and increases while control signal CS0 is high. When control signal CS0 is low, the inductor current iL decreases and, thus, the sense current iRsense—0 decreases, until the control signal CS0 is high again. The signal graphs 200 depict operation of switching power converter 102 in continuous conduction mode. In continuous conduction mode, the sense current iRsense—0 is always either increasing or decreasing. Thus, the sense resistor 116 (FIG. 1) conducts current for the entire period of control signal CS0 in continuous conduction mode.
For the low RMS input voltage VX—LOW RMS, the duty cycle of control signal CS0 is larger than the duty cycle of the high RMS input voltage VX—HIGH RMS because more current is needed by load 112 to supply the power demand of load 112. (“Duty cycle” is the ratio of the high time of control signal CS0 to the period of control signal CS0.) Because the sense current iRsense—0 continues to increase when control signal CS0 is high, a larger duty cycle of control signal CS0 results in a larger average sense current iRsense—0 for the low RMS input voltage VX—LOW RMS relative to the high RMS input voltage VX—HIGH RMS. Consequently, the combination of a high sense current iRsense—0 and a high duty cycle result in a large power dissipation in the current sense resistor 116 during the low RMS input voltage VX—LOW RMS. Thus, the largest power dissipation occurs during the low RMS input voltage VX—LOW RMS when efficiency of the switching power converter 102 is generally lower. In at least one embodiment, power losses are a combination of i2R losses for resistive elements, switching losses proportional to iL·VX, and capacitive losses proportional to CV2. In at least one embodiment, low RMS input voltage VX—LOW RMS represent the highest current iL and the highest i2R losses. In at least one embodiment, the other losses do not change or the rise is negligible compared to the i2R loss increase. Thus, in at least one embodiment, the i2R losses either dominate or rise faster than other losses fall.
Referring to FIGS. 3 and 4, power control system 300 is identical to power control system 100 except the current sense resistor 116 is replaced with a current sense resistor 302. FIG. 4 depicts signal graphs 300 with power control system 300 operating in continuous conduction mode. Current sense resistor 302 is connected in series with switch 108 and conducts sense current iRsense—1. Thus, sense current iRsense—1 is zero when switch 108 is nonconductive, and sense current iRsense—1 increases as inductor current iL increases when switch 108 conducts. As discussed in conjunction with FIG. 2, the duty cycle of control signal CS1 is high during low RMS voltage VX—LOW RMS and low during high RMS voltage VX—HIGH RMS. Thus, the average sense current iRsense—1 is higher during low RMS voltage VX—LOW RMS than during high RMS voltage VX—HIGH RMS. Power control system 300 reduces the overall power dissipation of current sense resistor 302 versus the power dissipation of current resistor 116 (FIG. 1) since current sense resistor 302 only dissipates power when control signal CS1 is high. However, as with power control system 100, the largest power dissipation occurs during the low RMS input voltage VX—LOW RMS when efficiency of the switching power converter 102 is generally lower.